Channel impulse response estimating decision feedback equalizer

ABSTRACT

A decision feedback equalizer is operated by making first symbol decisions from an output of the decision feedback equalizer such that the first symbol decisions are characterized by a relatively long processing delay, by making second symbol decisions from the output of the decision feedback equalizer such that the second symbol decisions are characterized by a relatively short processing delay, and by determining tap weights for the decision feedback equalizer based on the first and second symbol decisions. The first symbol decisions may be derived from the output of a long traceback trellis decoder. The second symbol decisions may be derived either from the output of a short traceback trellis decoder or from shorter delay outputs of the long traceback trellis decoder.

RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Application60/571,447 filed on May 14, 2004.

TECHNICAL FIELD OF THE INVENTION

The present invention relates to the estimation of channel impulseresponses for decision feedback equalizers.

BACKGROUND OF THE INVENTION

Since the adoption of the Advanced Television Systems Committee (ATSC)digital television (DTV) standard in 1996, there has been an ongoingeffort to improve the design of receivers built for the ATSC DTV signal.The primary obstacle that faces designers in designing receivers so thatthey achieve good reception is the presence of multipath interference inthe broadcast television channel.

The broadcast television channel is a relatively severe multipathenvironment due to a variety of conditions that are encountered in thechannel and at the receiver. Strong interfering signals may arrive atthe receiver both before and after the largest amplitude signal. Inaddition, the signal transmitted through the channel is subject to timevarying channel conditions due to the movement of the transmitter andsignal reflectors, airplane flutter, and, for indoor reception, peoplewalking around the room. If mobile reception is desired, movement of thereceiver must also be considered. Designers add equalizers to receiversin order to cancel the effects of multipath interference and therebyimprove signal reception.

Because the channel is not known a priori at the receiver, the equalizermust be able to adapt its response to the channel conditions that itencounters and to changes in those channel conditions. To aid in theconvergence of an adaptive equalizer to the channel conditions, thefield sync segment of the frame as defined in the ATSC standard may beused as a training sequence for the equalizer.

The frame as defined in the ATSC standard is shown in FIG. 1. Each framecontains two data fields, each data field contains 313 segments, andeach segment contains 832 symbols. The first four of these symbols ineach segment are segment sync symbols having the predefined symbolsequence [+5, −5, −5, +5].

The first segment in each field is a field sync segment. As shown inFIG. 2, the field sync segment comprises the four segment sync symbolsdiscussed above followed by a pseudo-noise sequence having a length of511 symbols (PN511) followed in turn by three pseudo-noise sequenceseach having a length of 63 symbols (PN63). Like the segment syncsymbols, all four of the pseudo-noise sequences are composed of symbolsfrom the predefined symbol set {+5, −5}. In alternate fields, the threePN63 sequences are identical; in the remaining fields, the center PN63sequence is inverted. The pseudo-noise sequences are followed by 128symbols, which are composed of various mode, reserved, and precodesymbols. The next 312 segments of the field are each comprised of thefour segment sync symbols followed by 828 8 level symbols that have beenencoded with a 12 phase trellis coder.

Because the first 704 symbols of each field sync segment are known,these symbols, as discussed above, may be used as a training sequencefor an adaptive equalizer. The original Grand Alliance receiver used anadaptive decision feedback equalizer (DFE) with 256 taps. The adaptivedecision feedback equalizer was adapted to the channel using a standardleast mean square (LMS) algorithm, and was trained with the field syncsegment of the transmitted frame.

However, because the field sync segment is transmitted relativelyinfrequently (about every 260,000 symbols), the total convergence timeof this equalizer is quite long if the equalizer adapts only on trainingsymbols prior to convergence. Therefore, it is known to use the symboldecisions made by the receiver in order to adapt equalizers to followchannel variations that occur between training sequences.

An adaptive decision feedback equalizer in an 8 VSB receiver would beexpected to use an 8 level slicer to make the symbol decisions thatwould be used to adapt the equalizer to the channel betweentransmissions of the training sequence. However, use of a symbol slicerresults in many symbol decision errors being fed to the feedback filterof the decision feedback equalizer when the channel has significantmultipath distortion or a low signal to noise ratio. These errors giverise to further errors resulting in what is called error propagationwithin the decision feedback equalizer. Error propagation greatlydegrades the performance of the decision feedback equalizer.

The present invention instead relies on decoders to avoid theconvergence and tracking problems of previous decision feedbackequalizers.

SUMMARY OF THE INVENTION

In accordance with one aspect of the present invention, a method ofoperating a decision feedback equalizer comprises the following: makingfirst symbol decisions from an output of the decision feedbackequalizer, wherein the first symbol decisions are characterized by arelatively long processing delay; making second symbol decisions fromthe output of the decision feedback equalizer, wherein the second symboldecisions are characterized by a relatively short processing delay; and,determining tap weights for the decision feedback equalizer based on thefirst and second symbol decisions.

In accordance with another aspect of the present invention, a decisionfeedback equalizer comprises a feed forward filter, a feedback filter, asummer, first and second decoders, and a tap weight controller. The feedforward filter receives data to be equalized. The summer combinesoutputs from the feed forward filter and the feedback filter to providean equalizer output. The first decoder is characterized by a relativelyshort processing delay, and the first decoder decodes the equalizeroutput to provide a first decoded equalizer output and supplies thefirst decoded equalizer output as an input to the feedback filter. Thesecond decoder is characterized by a relatively long processing delay,and the second decoder decodes the equalizer output to provide a seconddecoded equalizer output. The tap weight controller determines tapweights based on the first and second decoded equalizer outputs andsupplies the tap weights to the feed forward filter and the feedbackfilter.

In accordance with still another aspect of the present invention, adecision feedback equalizer comprises a feed forward filter, a feedbackfilter, a summer, first and second decoders, and a tap weightcontroller. The feed forward filter receives data to be equalized. Thesummer combines outputs from the feed forward filter and the feedbackfilter to provide an equalizer output. The first decoder decodes theequalizer output to provide a first decoded equalizer output andsupplies the first decoded equalizer output as an input to the feedbackfilter. The second decoder characterized by a relatively long processingdelay and by relatively shorter processing delays, the second decoderdecodes the equalizer output to provide a second decoded equalizeroutput in accordance with the relatively long processing delay and thirddecoded equalizer outputs in accordance with the relatively shorterprocessing delays. The tap weight controller determines tap weightsbased on the second and third decoded equalizer outputs and supplies thetap weights to the feed forward filter and the feedback filter.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features and advantages will become more apparent from adetailed consideration of the invention when taken in conjunction withthe drawings in which:

FIG. 1 illustrates a data frame according to the ATSC DTV standard;

FIG. 2 illustrates the field sync segment of the fields comprising thedata frame of FIG. 1;

FIG. 3 illustrates a tracking decision feedback equalizer systemaccording to embodiments of the present invention;

FIG. 4 is a timing diagram illustrating the non-zero time periodrequired for the calculation of a channel impulse estimate and updatedtap weights;

FIG. 5 is a timing diagram useful in illustrating a first method forimproving performance of a decision feedback equalizer in the presenceof time varying channel impulse responses; and,

FIG. 6 is a timing diagram useful in illustrating a second method forimproving performance of a decision feedback equalizer in the presenceof time varying channel impulse responses.

DETAILED DESCRIPTION

FIG. 3 illustrates a decision feedback equalizer system 10 that avoidsand/or mitigates the convergence and/or tracking problems of previousdecision feedback equalizers. The tap weights are calculated based onestimates of the channel impulse response. This arrangement makes use oftwo decoders, e.g., a short traceback trellis decoder 12 and a longtraceback trellis decoder 14. The short traceback trellis decoder 12,for example, may be a zero delay trellis decoder having a tracebackdepth of one, and the long traceback trellis decoder 14 has a longtraceback depth, such as a traceback depth of 32. Theses trellisdecoders are 12 phase trellis decoders with a delay equal to 12×(traceback depth−1).

The signal from the channel is processed by an automatic gain controller16, which provides the equalizer input signal y. A channel impulseresponse and noise estimator 18 uses the transmitted training sequenceas received in the equalizer input signal y and a stored version of thetransmitted training sequence to provide an estimate ĥ₀ of the channelimpulse response. A tap weight calculator 20 calculates a set of tapweights based on the initial estimate ĥ₀ of the channel impulse responseusing, for example, a minimum mean square error (MMSE) based algorithm,and supplies this set of tap weights to a decision feedback equalizer 22that includes a feed forward filter 24 and a feedback filter 26.

The decision feedback equalizer 22 equalizes the data symbols containedin the equalizer input signal y based on these training sequence basedtap weights and includes a summer 28 which supplies the output of thedecision feedback equalizer 22 to the short traceback trellis decoder 12and to the long traceback trellis decoder 14. The output, such as themaximum delay output, of the long traceback trellis decoder 14 forms thesymbol decisions b. The feedback filter 26 filters the output of theshort traceback trellis decoder 12, and the filtered output of thefeedback filter 26 is subtracted by the summer 28 from the output of thefeed forward filter 24 to provide the equalizer output.

The equalizer input signal y is delayed by a delay 30, and the delayedequalizer input signal y and the symbol decisions b are processed by aleast squares channel impulse and noise update estimator 32 thatproduces an updated channel impulse estimate ĥ_(LS). A tap weightcalculator 34 uses the updated channel impulse estimate ĥ_(LS) tocalculate an updated set of tap weights for the decision feedbackequalizer 22. The tap weights determined by the tap weight calculator 34are provided to the decision feedback equalizer 22 during periods whenthe tap weights based on the training sequence are not available fromthe tap weight calculator 20. The delay imposed by the delay 30 is equalto the combined processing delay of the decision feedback equalizer 22and the long traceback trellis decoder 14.

Because the data symbols in an 8-VSB system are trellis coded, it isdesirable to make use of the long traceback trellis decoder 14 as thesymbol decision device to supply symbol decisions to the least squarechannel impulse and noise update estimator 32. By using a trellisdecoder instead of a symbol slicer, the number of symbol decision errorssupplied to the feedback filter 26 is reduced.

The reliability of a trellis decoder is proportional to its tracebackdepth. The long traceback trellis decoder 14, because of its longertraceback depth, produces more reliable decisions. However, because ofits longer traceback depth, the decision process of the long tracebacktrellis decoder 14 incurs a longer delay.

By contrast, the symbol decisions of the short traceback trellis decoder12 are less reliable because of its shorter traceback depth. However,while its symbol decisions are less reliable than a trellis decoder witha longer delay, the short traceback trellis decoder 12 is stillsignificantly more reliable than an 8 level symbol slicer.

It is well known that a symbol decision device with a delay greater thanzero creates a problem for a decision feedback equalizer with respect tothe cancellation of short delay multipath. Therefore, decision feedbackequalizers for 8 VSB receivers with a zero delay trellis decoder in thefeedback loop have been used to reduce error propagation. Thus, thedecision feedback equalizer 22 uses the short traceback trellis decoder12 in the feedback loop of the feedback filter 26.

As indicated above, the output of the decision feedback equalizer 22 isthe output of the summer 28. This output is fed to the long tracebacktrellis decoder 14. The long traceback trellis decoder 14 has a longtraceback depth (e.g., traceback depth=32, delay=12×31=372 symbols). Thelong traceback trellis decoder 14 provides the final bit decisions forthe subsequent stages of the receiver in which the decision feedbackequalizer 22 is used. Also, as described below, the long tracebacktrellis decoder 14 provides symbol decisions used by the least squareschannel impulse and noise update estimator 32 in order to calculate theupdated channel impulse response estimate that is used by the tap weightcalculator 34 to calculate the updated tap weights for the decisionfeedback equalizer 22 so that the decision feedback equalizer 22 canfollow variations in the channel impulse response that occur betweentraining sequences.

Thus, the channel impulse response estimate ĥ₀ is formed by the channelimpulse response and noise estimator 18 from the received trainingsequence and a set of tap weights are calculated by the tap weightcalculator 20 from that channel impulse response estimate. Then, as thedecision feedback equalizer 22 runs, reliable symbols decisions aretaken from the long traceback trellis decoder 14 as relatively longpseudo training sequences, and these relatively long pseudo trainingsequences are used by the least squares channel impulse and noise updateestimator 32 to calculate the updated channel impulse response estimatesĥ_(LS) from which updated decision feedback equalizer tap weights arecalculated by the tap weight calculator 34. This process allows for thetracking of time varying channel impulse responses.

As indicated above, the channel impulse response estimate ĥ₀ is based onthe received training sequence. The channel impulse response estimate ĥ₀to be estimated is of length L_(h)=L_(ha)+L_(hc)+1 where L_(ha) is thelength of the anti-causal part of the channel impulse response estimateĥ₀ and L_(hc) is the length of the causal part of the channel impulseresponse estimate ĥ₀. The length of the training sequence is L_(n).

It may be assumed that the L_(n) long vector of the a priori knowntraining symbols as transmitted is given by the following expression:a=[a ₀, - - - , a _(L) _(n) ⁻¹]^(T)  (1)The vector of received symbols is given by the following equation:y=[y _(L) _(hc) , - - - , y _(L) _(n) _(−L) _(ha) ⁻¹]^(T)  (2)The first received training data element is designated y₀. Typically,this would mean that y₀ contains a contribution from the firsttransmitted training symbol multiplied by the maximum magnitude tap ofthe channel impulse response vector h. Note that the vector y containsdata elements comprised of contributions due to multipath only of apriori known training symbols. Also, the vector y does not include y₀which may contain contributions of unknown 8 level symbols due tomultipath.

A convolution matrix A of size (L_(n)−L_(ha)−L_(hc))×(L_(ha)+L_(hc)+1)may be formed from the known transmitted training symbols as given bythe following equation:

$\begin{matrix}{A = \begin{bmatrix}a_{{Lha} + {Lhc}} & a_{{Lha} + {Lhc} - 1} & - & - & - & a_{0} \\a_{{Lha} + {Lhc} + 1} & a_{{Lha} + {Lhc}} & \; & \; & \; & a_{1} \\ - & - & \; & \; & \; & - \\ - & - & \; & \; & \; & - \\ - & - & \; & \; & \; & - \\a_{{Ln} - 1} & a_{{Ln} - 2} & - & - & - & a_{{Ln} - {Lha} - {Lhc} - 1}\end{bmatrix}} & (3)\end{matrix}$Because the vector y of received symbols is given by the followingequation:y=Ah+v  (4)where h is the channel impulse response vector of length L_(h) and v isa noise vector, the least squares channel impulse response estimate isgiven by the solution of equation (4) according to the followingequation:ĥ ₀=(A ^(T) A)⁻¹ A ^(T) y  (5)However, this method is only effective if L_(n) satisfies the followinginequality:L _(n)≧2(L _(ha) +L _(hc))−1  (6)If the training sequence is too short with respect to the length of thechannel impulse response, then this method does not produce a goodresult because the system of equations given by equation (4) to besolved is underdetermined, which is often the case for 8 VSB terrestrialchannels. For example, with L_(n)=704, the channel impulse response mustbe less than 352 symbols long. However, longer channel impulse responsesare commonly found in practice.

A better method for finding the channel impulse response is based on amodified form of the convolution matrix A. A long vector a of lengthL_(n) of a priori known training symbols is again given by theexpression (1). However, the convolution matrix A this time is an(L_(n)+L_(ha)+L_(hc))×L_(h) convolution matrix comprising trainingsymbols and zeros and is given by the following equation:

$\begin{matrix}{A = \begin{bmatrix}a_{0} & \; & 0 & - & - & - & - & - & - & \; & 0 \\ - & \; & a_{0} & 0 & \; & \; & \; & \; & \; & \; & - \\ - & \; & \; & \; & \; & \; & \; & \; & \; & \; & - \\ - & \; & \; & \; & \; & \; & \; & \; & 0 & \; & 0 \\a_{{Lh} - 2} & \; & - & - & - & - & - & - & a_{0} & \; & 0 \\a_{{Lh} - 1} & \; & - & - & - & - & - & - & - & \; & a_{0} \\ - & \; & \; & \; & \; & \; & \; & \; & \; & \; & - \\ - & \; & \; & \; & \; & \; & \; & \; & \; & \; & - \\ - & \; & \; & \; & \; & \; & \; & \; & \; & \; & - \\a_{{Ln} - 1} & \; & - & - & - & - & - & - & - & \; & a_{{Ln} - {Lh}} \\0 & \; & a_{{Ln} - 1} & \; & \; & \; & \; & \; & \; & \; & a_{{Ln} - {Lh} - 1} \\ - & \; & 0 & \; & \; & \; & \; & \; & \; & \; & \; \\ - & \; & \; & \; & \; & \; & \; & \; & \; & \; & \; \\{- \;} & \; & \; & \; & \; & \; & \; & \; & a_{{Ln} - 1} & \; & a_{{Ln} - 2} \\0 & \; & - & - & - & - & - & - & 0 & \; & a_{{Ln} - 1}\end{bmatrix}} & (7)\end{matrix}$

The vector of received symbols is given by the following equation:y=[y _(−Lha), - - - , y ₀, - - - , y _(Ln+Lhc−1)]^(T)  (8)where y₀ through y_(Ln−1) are the received training symbols. So, thevector of equation (8) contains the known training symbols as well ascontributions from random symbols before and after the training sequencedue to multipath.

Again, equation (4) needs to be solved. Now, the convolution matrix A isa taller matrix because zeros have been substituted for the unknownsymbols that surround the training sequence. This new convolution matrixA yields an over-determined system of equations.

The initial channel impulse response and noise estimator 18 solvesequation (4) according to equation (5) using the new convolution matrixA of equation (7) and vector y of equation (8) to produce the channelimpulse response estimate ĥ₀. More complicated methods may be utilizedto give even more accurate results, if necessary.

The tap weight calculator 20 uses the channel impulse response estimateĥ₀ to calculate a set of minimum mean square error (MMSE) tap weightsfor the decision feedback equalizer 22. Methods for calculating minimummean square error tap weights from a channel impulse response are wellknown. Alternatively, the tap weight calculator 20 may use other methodssuch as the zero-forcing method to calculate the tap weights.

Accurate channel impulse response estimate updates can also becalculated between training sequences (when only a priori unknownsymbols are received). For example, a least squares channel impulseresponse estimation may be calculated from an over determined system ofequations. Dynamic changes to the channel impulse response may beaccurately tracked by using receiver trellis decoder decisions on inputsymbols to form a long sequence of near perfectly decoded symbols. Thissequence should have relatively few errors, even near threshold, and isselected to be long enough so that the underdetermined system problem ofthe “too short” 8 VSB training sequence is eliminated. The channelimpulse response may be, for example, updated as often as once persegment (or more or less often).

The updated channel impulse response to be estimated is, as before, oflength L_(h)=L_(ha)+L_(hc)+1 where L_(ha) is the length of theanti-causal part of the channel impulse response and L_(hc) is thelength of the causal part of the channel impulse response. A vector b oflength L_(b) is defined as the reliable trellis decoder decisions on theinput symbols that are provided by the long traceback trellis decoder14. Also, a Toeplitz matrix B is then defined according to the followingequation:

$\begin{matrix}{B = \begin{bmatrix}b_{{Lh} - 1} & b_{{Lh} - 2} & - & - & - & - & - & \; & b_{0} \\ - & b_{{Lh} - 1} & - & - & - & - & - & \; & - \\ - & - & \; & \; & \; & \; & - & \; & - \\ - & - & \; & \; & \; & \; & - & \; & - \\ - & - & \; & \; & \; & \; & - & \; & b_{{Lh} - 1} \\ - & - & \; & \; & \; & \; & - & \; & - \\ - & - & \; & \; & \; & \; & b_{{Lb} - {Lh}} & \; & - \\b_{{Lb} - 1} & b_{{Lb} - 2} & - & - & - & - & - & \; & b_{{Lb} - {Lh}}\end{bmatrix}} & (9)\end{matrix}$where the elements are real and consist of the symbol decisions in thevector b. To ensure an over determined system of equations, L_(b) isgiven by the following inequality:L _(b)≧2L _(h)−1  (10)The Toeplitz matrix B is of dimension (L_(b)−L_(h)+1)×L_(h) with(L_(b)−L_(h)+1)≧L_(h).

The received signal vector y has elements y_(i) forL_(hc)≦i≦(L_(b)−L_(ha)−1) where y_(i) is the received symbolcorresponding to the symbol decision b_(i). The received signal vector yis given by the following equation:y=Bh+v  (11)where h is the L_(h) long channel impulse response vector and v is anoise vector. The least squares solution for h is given by the followingequation:ĥ _(LS)=(B ^(T) B)⁻¹ B ^(T) y  (12)By utilizing reliable trellis decoder input symbol decisions, there issufficient support for calculating a channel impulse response estimatewith the required delay spread. As required by inequality (10), thevector b of symbol decisions must be at least twice as long as thechannel impulse response being estimated. The system of equations issufficiently over determined in order to diminish the adverse effect ofadditive White Gaussian Noise (AWGN). Therefore, a vector b of symboldecisions that is longer than twice the channel impulse response lengthis preferred.

The tap weight calculations performed by the tap weight calculator 20and the tap weight calculator 34 require not only a channel impulseresponse estimate but also a noise estimate. The noise may be estimatedby calculating an estimate of the received vector y according to ŷ=Aĥwhere ĥ is the latest calculated channel impulse response estimate.Then, the noise estimation is given by the following equation:

$\begin{matrix}{{\hat{\sigma}}^{2} = \frac{{{\hat{y} - y}}^{2}}{{length}(y)}} & (13)\end{matrix}$where ∥. ∥ is the 2-norm.

In order to apply the above equations to an 8 VSB receiver, thefollowing parameters may be used as an example: L_(h)=512, L_(ha)=63,L_(hc)=448, L_(b)=2496, and L_(n)=704. The vector b is formed from asequence of trellis decoder decisions made by the long traceback trellisdecoder 14 on the input symbols. The delay (31×12=372) of the longtraceback trellis decoder 14 is not significant compared to a channelimpulse response estimate update rate of once per segment. Normally, thelong traceback trellis decoder 14 would just make output bit pairdecisions, but it can also make equally reliable decisions on the inputsymbols.

The vector b, for example, may be selected as three segments (L_(b)=2496symbols) long. So, three data segments may be used to produce a singlechannel impulse response estimate update. A new channel impulse responseupdate can be obtained once per segment by proceeding in a slidingwindow manner. Optionally, several consecutive channel impulse responseestimate updates can be averaged in order to further improve channelimpulse response accuracy if necessary. This additional averaging can bea problem if the channel impulse response is varying rapidly.

A vector b with fewer than three segments of symbol decisions may beused provided that, as stated in inequality (10), the length of thevector b is at least twice as long as the channel impulse response to beestimated. As previously stated, however, long b vectors help todiminish the adverse effects of AWGN.

The latency time (which may be referred to as Tap Update Latency or TUL)involved in updating the decision feedback equalizer 22 with new tapweights is caused by the sum of (i) the symbol decision delay of thelong traceback trellis decoder 14, (ii) the time delay resulting fromthe calculation by the least squares channel impulse and noise updateestimator 32 of the channel impulse response estimate update, and (iii)the time delay resulting from the calculation by the tap weightcalculator 34 of the MMSE tap weights.

The delay of the first item (i) may be reduced if, instead of using onlydecisions of the long traceback trellis decoder 14 for the channelimpulse response estimate update, a combination of symbol decisions fromthe long traceback trellis decoder 14 and the short traceback trellisdecoder 12 is used. The use of this combination of symbol decisions isillustrated in FIGS. 4 and 5.

The first row of the timing diagram in FIG. 4 represents a series ofsegment time periods containing corresponding segments of receivedsymbols y as they are input to the decision feedback equalizer 22.

The second row represents the delay that the processing of the decisionfeedback equalizer 22 imposes on these segment time periods as thecorresponding equalized segments exit from the output of the decisionfeedback equalizer 22 and are provided to the long traceback trellisdecoder 14. As shown in FIG. 4, the processing of the decision feedbackequalizer 22 delays the segments in time relative to the correspondingsegments at the input of the decision feedback equalizer 22.

The third row represents the additional delay that the processing of thelong traceback trellis decoder 14 imposes on these segment time periodsas the corresponding segments of symbol decisions exit from the outputof the long traceback trellis decoder 14 and are provided to the leastsquares channel impulse and noise update estimator 32. As shown in FIG.4, the processing of the long traceback trellis decoder 14 delays thesymbol decisions in time relative to the corresponding equalizedsegments (second row) at the input of the long traceback trellis decoder14.

The fourth row represents the additional delays of making the channelimpulse response and tap weight calculations by the least squareschannel impulse and noise update estimator 32 and the tap weightcalculator 34. For the sake of convenience (and not of necessity), itmay be assumed that each of the delays given in items (i), (ii), and(iii) above is a ½ segment delay. With these assumptions, the updatedtap weights calculated by the tap weight calculator 34 from the vector bthat is composed of the symbol decisions in the three segment timeperiods 1, 2, and 3 will not be applied to the decision feedbackequalizer 22 until after the second half of the equalized segment insegment time period 5 begins being output from the decision feedbackequalizer 22. This corresponds to a 1.5 segment update delay.Accordingly, the tap update latency TUL is 1.5 segments.

In a channel whose channel impulse response is rapidly changing, thisdelay between (i) the time that segments are processed by the decisionfeedback equalizer 22 and (ii) the time at which the updated tap weightscalculated on the basis on these segments are applied to the decisionfeedback equalizer 22 may degrade performance of the decision feedbackequalizer 22 because the channel impulse response changes too muchbetween the end of segment 3 and the beginning of segment 5.

Several delay assumptions are made above for the purpose of a clearexplanation. However, these assumptions are not intended to be limiting.

The timing diagram of FIG. 5 shows an improved method of determining thetap weights to be supplied to the decision feedback equalizer 22. Here,2.5 segments of symbol decisions b from the long traceback trellisdecoder 14 plus 0.5 segments of symbol decisions c from the shorttraceback trellis decoder 12 are used by the least squares channelimpulse and noise update estimator 32 to form a three segment longdecision vector b that it then uses to produce the updated channelimpulse estimate ĥ_(LS).

The size of the portion of the three segment long decision vector b thatis contributed by the short traceback trellis decoder 12 is chosen to beequal to the delay imposed by the processing of the long tracebacktrellis decoder 14. As an example, given the assumptions discussedabove, this delay is 0.5 segment and removes the delay imposed by theprocessing of the long traceback trellis decoder 14 from the tap updatelatency TUL, thereby reducing it to one segment.

Accordingly, the first row of the timing diagram in FIG. 5 represents aseries of segment time periods containing corresponding segments ofreceived symbols y as they are input to the decision feedback equalizer22.

The second row represents the delay that the processing of the decisionfeedback equalizer 22 imposes on these segment time periods as thecorresponding equalized segments exit from the output of the decisionfeedback equalizer 22 and are provided to the long traceback trellisdecoder 14. As shown in FIG. 5, the processing of the decision feedbackequalizer 22 delays the segments in time relative to the correspondingsegments at the input of the decision feedback equalizer 22.

The third row represents the zero delay that the processing of the shorttraceback trellis decoder 12 imposes on these segment time periods asthe corresponding segments of symbol decisions exit from the output ofthe short traceback trellis decoder 12 and are provided to the leastsquares channel impulse and noise update estimator 32.

The fourth row represents the additional delay that the processing ofthe long traceback trellis decoder 14 imposes on these segment timeperiods as the corresponding segments of symbol decisions exit from theoutput of the long traceback trellis decoder 14 and are provided to theleast squares channel impulse and noise update estimator 32. As shown inFIG. 5, the processing of the long traceback trellis decoder 14 delaysthe symbol decisions in time relative to the corresponding equalizedsegments at the input of the long traceback trellis decoder 14.

The fifth row represents the additional delays of making the channelimpulse response and tap weight calculations by the least squareschannel impulse and noise update estimator 32 and the tap weightcalculator 34.

As shown in FIG. 5, the least squares channel impulse and noise updateestimator 32 uses 2.5 segments of symbol decisions b from the longtraceback trellis decoder 14 and 0.5 segments of symbol decisions c fromthe short traceback trellis decoder 12 in the calculation of the updatedchannel impulse estimate ĥ_(LS). Given the assumption that the delayimposed by the long traceback trellis decoder 14 is 0.5 segment, thenthe 0.5 segments of symbol decisions c contributed by the shorttraceback trellis decoder 12 occur contemporaneously with the last halfsegment of the 2.5 segments of symbol decisions b contributed by thelong traceback trellis decoder 14.

The symbol decisions c of the short traceback trellis decoder 12 aresomewhat less reliable than the symbol decisions b of the long tracebacktrellis decoder 14. However, when the channel impulse response ischanging rapidly, as is the case with mobile receivers, the reduced thetap update latency TUL is a worthwhile tradeoff against the lessaccurate symbol decisions c.

The long traceback trellis decoder 14 has the capability of outputting areliable decision after a delay D_(max) equal to its maximum tracebackdepth minus 1. It is well known that the path memories internal to along traceback trellis decoder simultaneously hold symbol decisions ofdelay zero up to delay D_(max). These symbol decisions can be output inparallel at any desired time as shown by US published patent applicationUS2002/0154248 A1. This published application describes the use of suchparallel outputs to feed decisions back into the feedback filter of adecision feedback equalizer. This operation is effectively parallelloading where, for each symbol update, a new set of decisions withdelays of zero up to delay D_(max) are simultaneously loaded into thefeedback filter.

This concept may be applied by the least squares channel impulse andnoise update estimator 32 in determining the updated channel impulseestimate ĥ_(LS). Instead of using a combination of symbol decisions fromlong traceback trellis decoder 14 and the short traceback trellisdecoder 12 as described in relation to FIG. 5, a sufficient number ofthe parallel survivor path memory outputs of the long traceback trellisdecoder 14 is used as shown in FIG. 6 for the last 0.5 segment ofdecisions b′ needed for the channel impulse response estimate update,assuming a 0.5 segment delay. This method provides the desired reductionin tap update latency TUL and, at the same time, uses more reliablesymbol decisions compared to using a combination of symbol decisionsfrom the long traceback trellis decoder 14 and the short tracebacktrellis decoder 12 as described in relation to FIG. 5.

Accordingly, the first row of the timing diagram in FIG. 6 represents aseries of segment time periods containing corresponding segments ofreceived symbols y as they are input to the decision feedback equalizer22.

The second row represents the delay that the processing of the decisionfeedback equalizer 22 imposes on these segment time periods as thecorresponding equalized segments exit from the output of the decisionfeedback equalizer 22 and are provided to the long traceback trellisdecoder 14. As shown in FIG. 6, the processing of the decision feedbackequalizer 22 delays the segments in time relative to the correspondingsegments at the input of the decision feedback equalizer 22.

The third row represents the zero to D_(max) delay that the processingof the long traceback trellis decoder 14 imposes on these segment timeperiods as the corresponding segments of symbol decisions are output inparallel from the long traceback trellis decoder 14 and are provided tothe least squares channel impulse and noise update estimator 32.

The fourth row represents the additional delay that the processing ofthe long traceback trellis decoder 14 imposes on these segment timeperiods as the corresponding segments of symbol decisions exit from theoutput of the long traceback trellis decoder 14 and are provided to theleast squares channel impulse and noise update estimator 32. As shown inFIG. 6, the processing of the long traceback trellis decoder 14 delaysthe symbol decisions in time relative to the corresponding equalizedsegments at the input of the long traceback trellis decoder 14.

The fifth row represents the additional delays of making the channelimpulse response and tap weight calculations by the least squareschannel impulse and noise update estimator 32 and the tap weightcalculator 34.

As shown in FIG. 6, the least squares channel impulse and noise updateestimator 32 uses 2.5 segments of symbol decisions b from the output ofthe long traceback trellis decoder 14 and 0.5 segments of parallelsymbol decisions b′ from the long traceback trellis decoder 14 in thecalculation of the updated channel impulse estimate ĥ_(LS). Given theassumption that the delay imposed by the long traceback trellis decoder14 is 0.5 segment, then the 0.5 segment of parallel symbol decisionscontributed by the long traceback trellis decoder 14 occurcontemporaneously with the last half segment of the 2.5 segments ofsymbol decisions contributed by the output of the long traceback trellisdecoder 14.

The parallel symbol decisions from the long traceback trellis decoder 14(see b′ in FIG. 3) are more reliable than the symbol decisions of theshort traceback trellis decoder 12.

Certain modifications of the present invention have been discussedabove. Other modifications of the present invention will occur to thosepracticing in the art of the present invention. For example, thedecoders 12 and 14 may be 12-phase trellis decoders. The use of 12-phasetrellis decoders is, for the most part, specific to the digitaltelevision application in compliance with the ATSC standard. For otherapplications, however, decoders other than 12-phase trellis decoders maybe used.

Accordingly, the description of the present invention is to be construedas illustrative only and is for the purpose of teaching those skilled inthe art the best mode of carrying out the invention. The details may bevaried substantially without departing from the spirit of the invention,and the exclusive use of all modifications which are within the scope ofthe appended claims is reserved.

1. A method performed by a decision feedback equalizer having a feed forward filter and a feedback filter, comprising: making first symbol decisions from an output of the decision feedback equalizer, wherein the first symbol decisions are characterized by a relatively long processing delay; making second symbol decisions from the output of the decision feedback equalizer, wherein the second symbol decisions are characterized by a relatively short processing delay; obtaining an estimated channel impulse response update directly from the first and second symbol decisions; determining updated tap weights for both the feed forward filter and the feedback filter of the decision feedback equalizer directly from the estimated channel impulse response update; wherein the making of first symbol decisions comprises making the first symbol decisions by use of a first device having a first processing delay, wherein the making of second symbol decisions comprises making the second symbol decisions by use of a second device having a second processing delay; wherein the second processing delay is shorter than the first processing delay; wherein the making of the first symbol decisions by use of a device comprises making the first symbol decisions by use of a long traceback trellis decoder; and wherein the making of the second symbol decisions by use of a second device comprises making the second symbol decisions by use of a short traceback trellis decoder.
 2. The method of claim 1, wherein the short traceback trellis decoder comprises a zero delay trellis decoder.
 3. The method of claim 1 wherein the determining of tap weights comprises determining the tap weights based on an amount of the second symbol decisions commensurate with the relatively long processing delay.
 4. A method performed by a decision feedback equalizer having a feed forward filter and a feedback filter, comprising: making first symbol decisions from an output of the decision feedback equalizer, wherein the first symbol decisions are characterized by a relatively long processing delay; making second symbol decisions from the output of the decision feedback equalizer, wherein the second symbol decisions are characterized by a relatively short processing delay; obtaining an estimated channel impulse response update directly from the first and second symbol decisions; determining updated tap weights for both the feed forward filter and the feedback filter of the decision feedback equalizer directly from the estimated channel impulse response update; wherein the making of first symbol decisions comprises making symbol decisions b by use of a device that imposes a plurality of sequential processing delays on the output of the decision feedback equalizer during the making of the symbol decisions b; wherein the making of second symbol decisions comprises using symbol decisions b′ from the device; wherein the symbol decisions b′ are characterized by processing delays that are shorter than processing delays characterizing the symbol decisions b; wherein the making of symbol decisions b by use of a device comprises making the symbol decisions b by use of a long traceback trellis decoder; and wherein the using of symbol decisions b′ from the device comprises using the symbol decisions b′ from the long traceback trellis decoder.
 5. A decision feedback equalizer, comprising: a feed forward filter connected to receive symbols to be equalized; a feedback filter; a summer connected to combine outputs from the feed forward filter and the feedback filter to provide equalized symbols; a channel impulse and noise update estimator to obtain an output channel impulse response update; a tap weight calculator; a first decoder characterized by a relatively short processing delay, said first decoder including a trellis decoder, wherein the first decoder decodes the equalized symbols to provide first symbol decisions, wherein the first decoder supplies the first symbol decisions as an input to the feedback filter, and wherein the first decoder supplies the first symbol decisions to a first input of the channel impulse and noise update estimator; a second decoder characterized by a relatively long processing delay, said second decoder including a trellis decoder, wherein the second decoder decodes the equalized symbols to provide second symbol decisions, and wherein the second decoder supplies the second symbol decisions to a second input of the channel impulse and noise update estimator; and, wherein the tap weight calculator determines tap weights directly from the channel impulse and noise update estimator and supplies the tap weights to both the feed forward filter and the feedback filter.
 6. The decision feedback equalizer of claim 5, wherein the relatively short processing delay comprises a zero delay.
 7. The decision feedback equalizer of claim 5, wherein the first decoder comprises a short traceback trellis decoder, and wherein the second decoder comprises a long traceback trellis decoder.
 8. The decision feedback equalizer of claim 5, wherein the tap weight controller determines the tap weights based on an amount of the first decoded equalizer output commensurate with the relatively long processing delay.
 9. The decision feedback equalizer of claim 5, wherein the tap weight controller determines the tap weights in response to the data and the first and second decoded equalizer outputs.
 10. A decision feedback equalizer, comprising: a feed forward filter connected to receive data to be equalized; a feedback filter; a summer connected to combine outputs from the feed forward filter and the feedback filter to provide an equalizer output; a first decoder connected to decode the equalizer output to provide a first decoded output and supplies the first decoded output as an input to the feedback filter; a second decoder characterized by a relatively long processing delay and by relatively shorter processing delays, wherein the second decoder decodes the equalizer output to provide a second decoded output in accordance with the relatively long processing delay and third decoded outputs in accordance with the relatively shorter processing delays; a channel impulse and noise update estimator to obtain a channel impulse response update directly from the second and third decoded outputs; and a tap weight controller connected to determine tap weights directly from outputs of the channel impulse and noise update estimator and supplies the tap weights to the feed forward filter and the feedback filter; wherein the first decoder comprises a short traceback trellis decoder, and wherein the second decoder comprises a long traceback trellis decoder.
 11. The decision feedback equalizer of claim 10, wherein the first decoder comprises a zero delay decoder.
 12. The decision feedback equalizer of claim 11, wherein the zero delay trellis decoder comprises a short traceback trellis decoder.
 13. The decision feedback equalizer of claim 10, wherein the tap weight controller determines the tap weights based on an amount of the third decoded equalizer outputs commensurate with the relatively long processing delay.
 14. The decision feedback equalizer of claim 10, wherein the tap weight controller determines the tap weights in response to the data and the second and third decoded equalizer outputs.
 15. The decision feedback equalizer of claim 10, wherein the second decoder provides the third decoded outputs from parallel outputs of the second decoder, and wherein the parallel outputs of the second decoder represent different delays. 